Circuit for reducing third order intermodulation distortion for a broadband RF amplifier

ABSTRACT

A distortion control circuit for selective modulation of an RF signal includes an input port for coupling with an RF signal source, such as a multifrequency CATV signal, an output port for coupling to an associated RF amplifier, and a pair of selectively biased diodes for generating new third order products from the multifrequency RF signal which are the same magnitude, but opposite in phase to the nonlinear products generated by the RF amplifier. Since both the original multifrequency RF input signal and the new generated products from the distortion control circuit are applied to the input of the RF amplifier, the nonlinear products from the distortion control circuit and the RF amplifier will be canceled and the output of the RF amplifier will comprise only the multifrequency RF signal.

BACKGROUND OF THE INVENTION

This invention relates generally to radio frequency (RF) amplification.More particularly, the invention relates to a system for reducing thirdorder intermodulation distortion in broadband CATV RF amplifiers.

Lowering distortion in RF power amplifier circuits without compromisingtheir transient response is an omnipresent problem. High frequencyamplification is widely used in communications and broadcasting and alsowhere high-speed switching is required for use in digitalinstrumentation. However, high frequency amplifier applications extendlinear operation into areas where parasitic effects of interelectrodecapacitance, wire inductance, stored charge and even operating frequencywavelength begin to adversely affect and dominate circuit behavior.

Minimizing distortion is particularly important when a series ofamplifiers is cascaded over a signal transmission path, such as a seriesof RF amplifiers in a CATV transmission network. Disposed throughout aCATV transmission system are RF amplifiers that periodically amplify thetransmitted signals to counteract cable attenuation and attenuationcaused by passive CATV components, such as signal splitters andequalizers. The RF amplifiers are also employed to maintain the desiredcarrier-to-noise ratio. Due to the number of RF amplifiers employed in agiven CATV transmission system, each RF amplifier must provide minimumdegradation to the transmitted signal.

In an ideal communication system it is preferable that the componentswhich comprise the system are linear. However, as a practical reality,there are many nonlinearities that are typically introduced by theelectronic components, such as RF amplifiers. The distortions created byan RF amplifier which are of primary concern are second (even) and third(odd) order harmonic distortions. Prior art amplifier designs haveattempted to ameliorate the effects of even order distortions byemploying push-pull amplifier topologies, since the maximum even ordercancellation occurs when the proper 180° phase relationship ismaintained over the entire bandwidth. This is achieved through equalgain in both push-pull halves by matching the operating characteristicsof the active devices.

However, odd-order distortion is difficult to remedy. Odd-orderdistortion characteristics of an amplifier are manifest as crossmodulation (X-mod) and composite triple beat (CTB) distortions on thesignal being amplified. These are two types of intermodulation (IM)distortion. X-mod occurs when the modulated contents of one channelbeing transmitted interferes with and becomes part of an adjacent ornon-adjacent channel. CTB results from the combination of threefrequencies of carriers occurring in the proximity of each carrier sincethe carriers are typically equally spaced across the frequencybandwidth. Of the two noted distortions, CTB becomes more problematicwhen increasing the number of channels on a given CATV system. WhileX-mod distortion also increases in proportion to the number of channels,the possibility of CTB is more dramatic due to the increased number ofavailable combinations from among the total number of transmittedchannels. As the number of channels transmitted by a communicationsystem increases, or as the channels reside closer together, theodd-order distortion becomes a limiting factor of amplifier performance.

The nonlinear properties of an RF amplifier can be described by a curvewhich can be expanded into a Taylor series as follows:

Uout=a ₁ ·U _(in) +a ₂ ·U _(in) ² +a ₃ ·U _(in) ³ +a ₄ ·U _(in) ⁴ +a ₅·U _(in) ⁵ . . . a _(n) ·U _(in) ^(n),  Equation 1

where U_(in) is the input potential and Uout is the output potential anda_(n) is a factor that determines the magnitude of the term. It shouldbe noted that a₁·U_(in) is the 1st order term; a₂·U_(in) ² is the 2ndorder term; a₃·U_(in) ³ is the 3rd order term . . . and a_(n)·U_(in)^(n) is the nth order term. The magnitudes of the individual terms arestrongly dependent on the input signal and, therefore, on the levelcontrol of the amplifier.

If we have the following:

U _(in) =A·cos ω_(i) t  Equation 2

There exists for each term multiple combination possibilities of theinput circulating frequencies ω_(i) due to the power of thecorresponding order number. Additionally, in multifrequency transmissionsystems, such as a CATV transmission network, the number of newcirculating frequencies at the output of the network increasesexponentially with the number (i) of frequencies at the input. These newcirculating frequencies are referred to herein as intermodulation (IM)products and are detrimental to the accurate acquisition of CATVsignals.

Using the Taylor series, it can be demonstrated that all odd order termscreate products which appear at the same location as the lower-valuedodd order terms. Therefore, the third order term creates a product atthe base frequency, (odd order number 1), the fifth order term creates aproduct at the third order and one at the base frequency. If the inputsignal consists, for example, of the base circulating frequencies ω₁ andω₂ (i=2) with the same amplitude A, that is expressed with:$\begin{matrix}\begin{matrix}{U_{in} = \quad {{A\quad \cos \quad \left( {\omega_{1}t} \right)} + {A\quad \cos \quad \left( {\omega_{2}t} \right)}}} \\{{= \quad {0.5{A\left( {^{j\quad \omega_{1}t} + ^{{- j}\quad \omega_{1}t} + ^{j\quad \omega_{2}t} + ^{{- j}\quad \omega_{2}t}} \right)}}},}\end{matrix} & {{Equation}\quad 3}\end{matrix}$

then the third order term a₃·U_(in) ³ of Equation 1, (which is ofinterest in the present application), creates the following newproducts: ±ω₁; ±ω₂; ±3ω₁; ±3ω₂; ±(2ω₂±ω₁); ±(2ω₁±ω₂). In this case thereare 16 new circulating output frequencies due to two input circulatingfrequencies.

A “weakly” nonlinear transmission system can be defined such that: a)the effect of odd order terms on other lower-valued odd order terms isnegligibly small; and b) higher-valued terms after the third order termare negligibly small. Accordingly, a weakly nonlinear system may bemathematically described such that the Taylor series is broken off afterthe third order term (a₃·U_(in) ³). Weakly nonlinear systems arecharacterized in that a 1 db increase in the level of the inputcirculating frequencies ω_(i) causes an increase of 3 db in the thirdorder IM products.

Communication systems, such as CATV systems which include broadband RFamplifiers, are further regarded as dynamically nonlinear systemswhereby the amplitudes and phases of the IM products are dependent uponthe input frequencies.

There are three basic ways of correcting distortion created by anon-linear device: 1) reduce the signal power level; 2) use a feedforward technique; and 3) use a predistortion or postdistortiontechnique. The first method reduces the signal power level such that thenon-linear device is operating in its linear region. In the case of anRF amplifier this results in very high power consumption for low RFoutput power. Of course, the high power consumption is a disadvantage.However, this method is not an option if high output level is requiredon a permanent basis.

The second method is the feed forward technique. Using this technique,the input signal of the main amplification circuit is sampled andcompared to the output signal to determine the difference between thesignals. This difference is the distortion component which is amplifiedby an auxiliary amplification circuit and combined with the output ofthe main amplification circuit such that the two distortion componentscancel each other. However, the power consumed by the auxiliaryamplification circuit is comparable to that consumed by the mainamplification circuit and the circuitry is also complex and expensive.At the upper frequency limit it is very difficult to maintain themagnitude and phase conditions with respect to temperature.

The third method is the pre- or post-distortion technique. Dependingupon whether the compensating distortion signal is generated before thenon-linear device or after, the respective term predistortion orpostdistortion is used. In this technique, a distortion signal equal inamplitude but opposite in phase to the distortion component generated bythe amplifier circuit is estimated and generated. This is used to cancelthe distortion at the input (for predistortion) or output (forpostdistortion) of the amplifier, thereby improving the operatingcharacteristics of the amplifier.

SUMMARY

The present invention is a distortion control circuit for selectivemodulation of an RF signal. The present invention includes an input portfor coupling with an RF signal source, such as a multifrequency CATVsignal, and an output port for coupling to an associated electricalcircuit such as a hybrid RF amplifier. Hybrid RF amplifiers are knownfor generating unwanted nonlinear characteristics, particularly secondand third order products. The present invention generates new thirdorder products from the multifrequency RF signal which are the samemagnitude, but opposite in phase to the nonlinear products generated bythe hybrid RF amplifier. Since both the original multifrequency RF inputsignal and the new generated products from the invention are coupled tothe input of the hybrid RF amplifier, the nonlinear products from thepresent invention and the hybrid RF amplifier will be canceled and theoutput of the hybrid RF amplifier will comprise only the multifrequencyRF signal. The distortion control circuit includes a nonlinear circuithaving a pair of diodes which are selectively biased to create adistortion product for adding to the input signal which is opposite inphase and equal in magnitude to any distortion created by the hybrid RFamplifier. The present inventive circuit is particularly adaptableweakly nonlinear systems and provides the ability to largely match thedynamically nonlinear behavior of a system to be compensated and achievecompensation over a frequency range of at least 860 MHz.

Accordingly, it is an object of the present invention to provide acircuit for reducing third order intermodulation distortion for abroadband RF amplifier.

Other objects and advantages of the system and the method will becomeapparent to those skilled in the art after reading a detaileddescription of the preferred embodiment.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of a circuit for reducing third orderintermodulation distortion in accordance with the present invention.

FIG. 2 is a graph of the diode differential forward resistance R_(F)verses the diode forward current I_(F).

FIG. 3 is a graph of the diode differential forward resistance R_(F)verses the diode forward current I_(F) with different resistors inseries with the diode.

FIG. 4 is an alternative embodiment of the present invention.

FIG. 5 is a bias control for the present invention.

DETAILED DESCRIPTION OF THE INVENTION

The preferred embodiment of the present invention will be described withreference to the drawing figures where like numerals represent likeelements throughout.

One basic structural element for a compensator is a nonlinear element.In accordance with the present invention, the nonlinear element ispreferably a Schottky diode pair. Diode current I_(F) and diode voltageU_(F) are generally related by the following equation:

I _(F) =I ₀(e ^(U) ^(_(F/m)) ^(·U) ^(_(T)) −1);  Equation 4

where: I_(F) is the diode forward current; I₀ is the diode inversecurrent; U_(F) is the diode forward voltage; m is a correction factorwith a value between 1 and 2; and U_(T) is the temperature dependentvoltage which can be written as: $\begin{matrix}{{U_{T} = \frac{k \cdot T}{e_{o}}};} & {{Equation}\quad 5}\end{matrix}$

whereby:

k=Boltznann's—constant (physical constant);

T=temperature in kelvin; and

e_(o)=electrical element charge (physical constant).

Accordingly, U_(T) is a constant for a certain temperature, (forexample, 25 mV at 23° C.). The Taylor series of an exponential functionyields a relatively large third order term: $\begin{matrix}{e^{x} = {1 + \frac{x}{\left( {1!} \right)} + \frac{x^{2}}{\left( {2!} \right)} + \frac{x^{3}}{\left( {3!} \right)}}} & {{Equation}\quad 6}\end{matrix}$

Equation 4 can be approximated by:

I _(F) ≈I _(o) e ^(kU) ^(_(F))   Equation 7

Which means that the diode forward current I_(F) is proportional to ane-function with the diode forward voltage U_(F) in its exponent. Sincethe diode is part of the inventive circuit, U_(F) is part of U_(in) andEquation 7 can be rewritten as:

I _(F) ≈I _(o) e ^(k·U) ^(_(in))   Equation 8

Assuming that k·U_(in) is x, and inserting it into Equation 6, the thirdorder term x³/3! will produce the same products, (i.e., compensatingproducts generated by the diode), as shown by Equation 3.

To achieve phase opposition of the compensating IM products relative tothose of the object to be compensated, the nonlinear element isconnected in accordance with the preferred embodiment of the presentinvention in the transverse branch of a T-member as shown in FIG. 1. Ifthe nonlinear element was arranged in the length branch of an equivalentT-member, the IM products would be in phase relative to those to becompensated, (provided the diodes in the two cases operate at the sameoperating points), and compensation would be impossible.

The system 10 for reducing third order IM in accordance with the presentinvention is located between an input E and an output A. The input Ecomprises a multifrequency operating signal, for example a CATV signalhaving plurality of CATV channels. The output A is connected to a systemto be compensated, for example a hybrid RF amplifier. As is well knownby those skilled in the art, the RF amplifier not only amplifies theoutput A, but also introduces undesired IM products. These IM productsare compensated for by the present invention. As shown, the system 10 isa circuit which comprises a plurality of resistors R₁, R₂, R₃, R₄; aplurality of capacitors C₁, C₂, C₃, C₄, C₅, C₆; and a nonlinear elementcomprising two Schottky barrier diodes D₁, D₂. As will be explained indetail hereinafter, the present invention produces third order“compensating” IM products, (including cross-modulation products), whichexhibit the same amplitude but opposite phase to the RFamplifier-generated IM products. The compensating IM products are addedat the node connecting the two resistors R₃ and R₄ and the two diodes D₁and D₂ such that the compensating IM products are added to themultifrequency operating signal and output at output A.

A control input S₁ is provided to control the operating point of thediodes D₁, D₂ and thereby control the magnitude of compensating IMproducts. At the control input S₁, a direct current (DC) is suppliedwhich flows through diodes D₁ and D₂ and determines the operating pointof the diodes D₁, D₂. A DC current change at the control input S₁influences the steepness of the diode characteristic. The change in thediode differential forward resistance RF in the lower segment of thediode characteristic is greater than in the upper segment, provided thatthe change of I_(F) is the same in both cases.

A graph of the diode differential forward resistance R_(F) verses thediode forward current I_(F) is shown in FIG. 2. It should be noted thatthis curve is an illustration of the frequency response at 10 KHz. Asshown, the diode differential forward resistance R_(F) is dependent uponboth the diode operating point and the change in diode forward currentI_(F). For example, the DC current at the control input S₁ changes theoperating point of the diodes D₁, D₂ from operating point A to operatingpoint B. At operating point A, a change in the diode forward currentI_(F) of 0.5 mA results in a change in the diode differential forwardresistance R_(F) of 155Ω. However, at operating point B, a change in thediode forward current I_(F) of 0.5 mA results in a change in the diodedifferential forward resistance R_(F) of only 2Ω. The differential diodecurrent ΔI_(F) is caused by the input level U_(in). Since U_(in) is anRF signal, it leads to an alternating diode current, as shown inEquation 8.

The magnitude of the compensating IM products is dependent upon thechange in diode differential forward resistance R_(F) as a function ofthe level of U_(in). Accordingly, a low DC current input at S₁ leads toa greater magnitude of compensating IM products, and a high DC currentinput at S₁ correspondingly leads to a smaller magnitude of compensatingIM products. By selectively controlling the DC current at input S₁, themagnitude of the compensating IM may be selectively controlled.

In accordance with the present invention, an additional method foradjusting the diode differential forward resistance R_(F) characteristicis provided by the introduction of the resistors R₁ and R₂, which froman AC standpoint are in series with D₁ and D₂. The signal currents areconducted from D₂ via R₂ directly to ground and from D₁ via R₁ and C₆ toground. C₆ is a blocking capacitor whose complex resistance isnegligibly small for all applied frequencies of the multifrequency inputsignal. The resistors R₁ and R₂ affect the diode differential forwardresistance R_(F) characteristic by flattening the characteristic,thereby influencing the magnitude of the compensating IM products.Referring to FIG. 3, a graph of the diode differential forwardresistance R_(F) verses the diode forward current I_(F) is shown. It canclearly be seen that as the value of the two resistors R₁, R₂ isincreased from 10Ω to 100Ω, the diode differential forward resistanceR_(F) characteristics changes from curve C to curve D.

Because of the circuit symmetry required to eliminate the second orderterms, the resistors R₁, R₂ have the same values and simultaneously playthe role of symmetry resistors, in that they equalize the potentiallydifferent diodes D₁, D₂. The symmetry resistors R₁ and R₂ decouple thetwo diodes D₁, D₂ from each other and lead to symmetrical compensatingIM products. In this manner, any deviations in the characteristics ofboth diodes D₁, D₂ are minimized.

The diode differential forward resistance R_(F) is frequency dependentdue to the complex parasitic effects of the diodes D₁, D₂. As the diodecurrent I_(F) decreases, the influence of the parasitic elementsincreases. The detrimental diffusion capacitance of the diodes D₁, D₂,(which parallels the diode differential forward resistance R_(F)),prevents the signal energy at the upper frequency limit from enteringinto the diode differential forward resistance R_(F). This leads to areduction of the compensating IM products at high frequencies.Unfortunately, a high level of compensating IM products is particularlyrequired at high frequencies because the IM products in RF amplifiersinherently increase with signal frequency due to the circuit-producedreverse coupling, which becomes less effective at high frequencies.

In order to counteract the parasitic effects, capacitors C₁, C₂ and C₃are provided. The common variable capacitor C₃ is used to match thefrequency distribution of the IM products of the RF amplifier, therebyachieving optimal broadband compensation. By providing the equalizingelement C₃, the required symmetry conditions can always be obtainedindependently of its adjustment. The capacitors C₄ and C₅ serveexclusively for blocking the direct components at the input E and outputA, and their capacitances are large enough such that they do not affectthe multifrequency operating signal. Together with R₁ and R₂, thecapacitors C₁, C₂ and C₃ provide frequency-dependent resistances, suchthat they permit a desired and controllable level of increase of thecompensating IM products at high frequencies. The flattening of thediode characteristic at high frequencies is counteracted by thereduction of the complex series resistances. As a result, the diodecharacteristic steepens not only at high frequencies, but an increase insteepness can also be realized if desired.

In accordance with the teachings of the present invention, animprovement of 8 db to 15 db in CTB and X-mod may be expected dependingupon the hybrid RF amplifier to which the system 10 is coupled.

An alternative embodiment 20 of the present invention is shown in FIG.4. In this alternative embodiment 20, C₃ is replaced by the variablecapacitance diode D₃. An input S₂ is provided for connection to avariable DC voltage. This permits electrical adjustments, for example,when the circuit 20 is integrated into a hybrid RF amplifier. The valueof R₅ is chosen to be very high so that there is no adverse influence tothe RF transmission behavior of the distortion system, (i.e., no adverseinfluence on insertion loss and return loss over the range of operatingfrequencies).

Referring to FIG. 5, a bias control 30 for the input S₁ of FIGS. 1 and 4is shown. The input voltage+U_(B) is the operating voltage which has afixed value. In the present case, the fixed value is 24 volts DC, whichcorresponds to the operating voltage of a hybrid RF amplifier. As thoseskilled in the art would appreciate, the value of +U_(B) may bedifferent in other applications. R₁₂ is a resistor with a positivetemperature coefficient to compensate temperature effects. It should benoted that although R₁₂ improves the temperature behavior, this resistoris optional. With the insertion of R₁₂ it is possible to realizetemperature compensation of the RF hybrid, (i.e., increased compensationas the temperature increases). Since R₁₃ is variable, it permitsadjustment of the compensation effect of the IM products.

Table 1 below sets forth the component values for the components shownin FIGS. 1, 4 and 5. It should be clearly recognized by those skilled inthe art that these component values have been selected for theparticular application and desired frequency range. These componentvalues are illustrative only and should not be considered to be anessential part of the present invention since they will change dependingupon the operating range of the system in which the distortion circuitis utilized and the amplifier to which the distortion circuit iscoupled. The values should not be viewed as limiting.

TABLE 1 Component Value C₁ 0.5 pF C₂ 0.5 pF C₃ 0.5-2 pF C₄, C₅, C₆ 1 nFR₁, R₂ 300 Ω to 750 Ω (dependent on application) R₃, R₄ 3.9 Ω R₅ 10 kΩ-100 k Ω D₁, _(D2) Schottky barrier diode pair D₃ Hyperabrupt variablecapacitance diode from 0.5-2.0 pF S₁ bias current 0, 8 mA to 2 mA R₁₁7.5 kΩ R₁₂ 2 kΩ R₁₃ 20 kΩ R₁₄ 7.5 kΩ

What is claimed is:
 1. A distortion control circuit for producing aselectively modulated signal comprising: a signal input port; anon-linear circuit coupled to said input port for selectively modulatinga signal received at said input port; said non-linear circuitcomprising: a pair of diodes coupled together in series; first andsecond capacitors coupled together in series, and coupled in parallel tosaid diode pair; a third capacitor, having a first end coupled betweensaid first and second coupled capacitors and a second end coupled toground; a first resistor coupled at one end to a first end of said diodepair and at a second end to a fourth capacitor, said fourth capacitorbeing further coupled to ground; and a second resistor coupled at oneend to a second end of said diode pair and at a second end to ground; athird resistor, coupled between said diodes and a fifth capacitor; andan output port coupled to said diode pair for outputting saidselectively modulated signal from said non-linear circuit; said fifthcapacitor coupled between said third resistor and said output port. 2.The circuit of claim 1, further including a first control signal inputport between said first resistor and said fourth capacitor.
 3. Thedistortion control circuit of claim 2 further including a low resistanceDC bias voltage circuit at said first control signal input port.
 4. Thecircuit of claim 1, wherein said third capacitor is a variablecapacitor.
 5. The circuit of claim 1, further including a fourthresistor, coupled between said diodes and a sixth capacitor, said sixthcapacitor being further coupled to said input port.
 6. The distortioncontrol circuit of claim 1, wherein said third capacitor is a variablecapacitance diode.
 7. The distortion control circuit of claim 6 furtherincluding a second control signal input port between said first andsecond coupled capacitors and said third capacitor.
 8. The distortioncontrol circuit of claim 7 further including a fifth resistor betweensaid second control signal input port and said first and second coupledcapacitors.
 9. A distortion control circuit for producing a selectivelymodulated signal comprising: a signal input port; a non-linear circuitcoupled to said input port for selectively modulating a signal receivedat said input port; said non-linear circuit comprising: a pair of diodescoupled together in series; first and second capacitors coupled togetherin series, and coupled in parallel to said diode pair; a thirdcapacitor, having a first end coupled between said first and secondcoupled capacitors and a second end coupled to ground; a first resistorcoupled at one end to a first end of said diode pair and at a second endto a fourth capacitor, said fourth capacitor being further coupled toground; and a second resistor coupled at one end to a second end of saiddiode pair and at a second end to ground; a low resistance DC biasvoltage circuit at a first control signal input port between said firstresistor and said fourth capacitor; a temperature compensation circuitcoupled with said bias circuit, for selectively adjusting said DC biasvoltage in response to a change in ambient temperature; and an outputport coupled to said diode pair for outputting said selectivelymodulated signal from said non-linear circuit.
 10. The distortioncontrol circuit of claim 9, whereby said bias circuit comprises at leastone resistor having a positive temperature coefficient.